1、<p><b>　　中文2180字</b></p><p>　　201x 屆 本 科 畢 業(yè) 設(shè) 計(jì)（外文翻譯）</p><p>　　學(xué) 院： </p><p>　　專(zhuān) 業(yè)： </p><p>　　姓 名：

2、 </p><p>　　學(xué) 號(hào)： </p><p>　　指導(dǎo)教師： </p><p>　　完成時(shí)間： </p><p><b>　　二〇一四年三月</b&g

3、t;</p><p>　　LTE的多址接入技術(shù)</p><p><b>　　LTE的多址接入</b></p><p><b>　　OFDM傳輸</b></p><p>　　正交頻分復(fù)用（OFDM）是一種多載波傳輸技術(shù)，已被采納為3gpplong長(zhǎng)期演化（LTE）的下行鏈路傳輸方案，也可用于其他幾個(gè)無(wú)

4、線技術(shù)，例如：wimax和DVB廣播技術(shù)。它的特點(diǎn)是在一個(gè)頻域內(nèi)分布著許多帶有間隔的子載波 △f=1/Tu其中， Tu是每個(gè)子載波的調(diào)制符號(hào)時(shí)間。如圖2-1所示，“OFDM子載波間隔 ”。</p><p>　　OFDM的傳輸是基于塊的。每個(gè)OFDM符號(hào)間隔之間，調(diào)制符號(hào)是并行發(fā)送的。調(diào)制符號(hào)可以通過(guò)調(diào)制字母表得到，如QPSK，16QAM或64QAM，對(duì)于3GPP組織LTE，子載波間隔是相等的為15 kHz。另一方

5、面，子載波的數(shù)目取決于傳輸帶寬，在一個(gè)10MHZ的頻譜分配下，600個(gè)子載波可以有序傳輸。當(dāng)然，帶寬減小了，子載波數(shù)目也相應(yīng)減少，帶寬增加了，子載波數(shù)目也相應(yīng)增加。</p><p>　　圖2-1 OFDM子載波間隔</p><p>　　在OFDM傳輸時(shí)，物理資源經(jīng)常被描述成一個(gè)時(shí)域—頻域的網(wǎng)格坐標(biāo)圖。在這個(gè)坐標(biāo)圖里一列對(duì)應(yīng)一個(gè)OFDM子載波，一行對(duì)應(yīng)一個(gè)OFDM子載波。如圖2-2所示，“

6、OFDM時(shí)頻網(wǎng)格” 。</p><p>　　盡管子載波的頻譜有重疊，但在理想情況下，是對(duì)OFDM子載波解調(diào)后不引起任何干擾的，這是因?yàn)閷?duì)每一個(gè)子載波間隔的特殊選擇，讓它等于相應(yīng)的解調(diào)符號(hào)率。</p><p>　　圖2-2 OFDM時(shí)頻網(wǎng)格</p><p>　　以一定的頻率fs= N ×△f進(jìn)行采樣的OFDM信號(hào)，是該size-N的逆離散傅立葉變換

7、（IDFT）的調(diào)制符號(hào)塊a0, a1,...aN-1。因此，OFDM調(diào)制可以通過(guò)IDFT處理再到數(shù)字-模擬的轉(zhuǎn)換來(lái)實(shí)現(xiàn)。（見(jiàn)圖2-3，“OFDM調(diào)制”）。在實(shí)際中，OFDM調(diào)制是以快速傅立葉反變換（IFFT）方式實(shí)現(xiàn)簡(jiǎn)單和快速的處理，通過(guò)選擇IDFT size N 等于2m（m為整數(shù)）。在接收端，對(duì)接收信號(hào)以fs= N ×△f的頻率采樣，高效的FFT處理是用來(lái)實(shí)現(xiàn)OFDM的解調(diào)和檢索調(diào)制符號(hào)塊a0, a1,...aN-1。（參

8、見(jiàn)圖2-4，“OFDM解調(diào)”）。</p><p>　　圖2-3 OFDM調(diào)制</p><p>　　圖2-4 OFDM解調(diào)</p><p>　　正如上面提到的，一個(gè)無(wú)干擾的OFDM信號(hào)可以解調(diào)出無(wú)任何子載波間干擾的信號(hào)。然而，在一個(gè)時(shí)間色散信道的情況下（如多徑無(wú)線信道），子載波之間的正交性丟失，造成符號(hào)間干擾（ISI）。這是因?yàn)?，解調(diào)器相關(guān)區(qū)間的一

9、條路徑將與不同路徑的符號(hào)邊界有重疊。（見(jiàn)圖2-5，“時(shí)間的分散性和相應(yīng)的接收信號(hào)”）。</p><p>　　圖2-5 次分散和相應(yīng)的接收信號(hào)</p><p>　　要解決這個(gè)問(wèn)題，使OFDM信號(hào)在無(wú)線信道傳播時(shí)對(duì)時(shí)間色散完全不敏感，所謂的插入循環(huán)前綴通常被使用。如圖2-6所示，“插入循環(huán)前綴”。循環(huán)前綴</p><p>　　插入就意味著OFDM符號(hào)的最后部分（第N個(gè)c

10、p）被復(fù)制并且被插入到OFDM塊的開(kāi)始部分。因此，OFDM符號(hào)的長(zhǎng)度從TU 到TU +TCP ,其中TCP =NCPTU是循環(huán)前綴的長(zhǎng)度。作為一個(gè)結(jié)果，OFDM符號(hào)率是減少的。因此，在時(shí)間色散信道里，只要時(shí)間色散的跨度小于循環(huán)前綴的長(zhǎng)度，子載波的正交性就能被保持。</p><p>　　圖2-6插入循環(huán)前綴</p><p>　　循環(huán)前綴插入的缺點(diǎn)是，在整個(gè)信號(hào)帶寬沒(méi)有減少，OFDM符號(hào)率減少

11、的情況下，就意味著在吞吐量方面有相應(yīng)的損失。OFDM調(diào)制組合（IFFT處理），一個(gè)（分散的）無(wú)線信道，以及解調(diào)（FFT處理）可以被看作是一個(gè)頻域信道。如圖2-7，“頻域模型的OFDM傳輸接收”，其中每個(gè)OFDM符號(hào)的時(shí)間期間，N個(gè)不同的調(diào)制碼元被發(fā)送，每一個(gè)在相應(yīng)的子載波上，在對(duì)比單一寬帶載波系統(tǒng)時(shí)，如WCDMAwhere，每個(gè)調(diào)制符號(hào)被傳輸在整個(gè)帶寬上。</p><p>　　圖2-7頻率的OFDM傳輸接收域模型

12、</p><p>　　在頻道k上，調(diào)制符號(hào)ak被縮放和相位轉(zhuǎn)移，通過(guò)復(fù)雜的信道系數(shù)Hk（頻域）。在接收端，解調(diào)后允許發(fā)送的信息準(zhǔn)確解碼。在接收端需要一個(gè)頻域的信道抽頭估計(jì)H0，H1， ...， HN-1。這可以通過(guò)在OFDM時(shí)頻網(wǎng)格內(nèi)以一定規(guī)律的間隔插入已知參考符號(hào)來(lái)實(shí)現(xiàn)，有時(shí)也稱(chēng)作導(dǎo)頻符號(hào)或?qū)ьl器。運(yùn)用參考符號(hào)的相關(guān)知識(shí)，接收機(jī)可以估計(jì)信道抽頭（頻域）用于解碼的必要。</p><p>&

13、lt;b>　　OFDM信號(hào)帶寬</b></p><p>　　一個(gè)OFDM信號(hào)的帶寬等于N×△f ，這就是說(shuō)：子載波數(shù)乘以子載波間隔數(shù)。另一方面，通過(guò)設(shè)置這個(gè)傳輸符號(hào)從一側(cè)組相鄰子載波到零，這個(gè)基帶被減少到NC×△f,其中NC 是非空子載波數(shù)目。然而，OFDM信號(hào)的頻譜脫落到基本帶寬以外的速度是很慢的，尤其比一個(gè)WCDMA信號(hào)慢的多。因此，在實(shí)際中，一個(gè)OFDM需要10%的保護(hù)

14、間隔。這也就是說(shuō)，舉個(gè)例子，在一個(gè) 5 MHZ 的頻譜分配中，OFDM基本帶寬 NC ×f 大約是4.5 MHZ。做一個(gè)假設(shè)，例如，為L(zhǎng)TE選擇一個(gè)15 KHZ的子載波間隔，那么，在 5MHZ內(nèi)應(yīng)對(duì)應(yīng)于300個(gè)子載波。</p><p>　　DFT OFDM傳輸</p><p>　　離散的傅里葉變換擴(kuò)展的正交頻分復(fù)用（DFTS-OFDM）已被用作LTE上行鏈路的傳輸方案。DFTS-

15、OFDM傳輸?shù)幕驹碓趫D2-8，“DFT的OFDM信號(hào)生成”中說(shuō)明。類(lèi)似于OFDM調(diào)制，DFTS-OFDM依賴(lài)于基于塊的信號(hào)生成。在DFTS-OFDM中，一個(gè)M調(diào)制符號(hào)塊來(lái)自于一些調(diào)制字母表，比如，QPSK 或者 16QAM,第一次被應(yīng)用到size-m DTF。這個(gè)DFT輸出被應(yīng)用到一個(gè)size-N 的逆DFT的連續(xù)輸入當(dāng)中。其中，N > M 且未使用的輸入（N-M）設(shè)置為零。和OFDM一樣，每個(gè)傳輸塊插入一個(gè)循環(huán)前綴。<

16、/p><p>　　圖2-8 DFT的OFDM信號(hào)的產(chǎn)生</p><p>　　與圖2-8，“DFT的OFDM信號(hào)生成”相比，基于IFFT OFDM調(diào)制的實(shí)現(xiàn)，很顯然，DFTS-OFDM可以看作是OFDM調(diào)制之前的DFT運(yùn)算。如果DFT的M的大小等于IDFT的N的大小，那么級(jí)聯(lián)DFT和IDFT的塊圖2-8“DFT的OFDM信號(hào)生成”將完全抵消。如果M小于N且IDFT的剩余輸入被設(shè)置為零，

17、則IDFT的輸出將是一個(gè)低功率變化的信號(hào)，類(lèi)似于一個(gè)單載波信號(hào)。此外，不同塊大小為m的瞬時(shí)帶寬發(fā)送的信號(hào)可以是多種多樣的，允許靈活的帶寬分配。</p><p>　　與DFTS-OFDM的主要好處想比，多載波傳輸方案，如OFDM,減少變化的瞬時(shí)發(fā)射功率，對(duì)提高功率放大器效率是可能的。功率的變化一般根據(jù)測(cè)得的峰值平均功率比（PRPA）來(lái)判斷。定義為在峰值功率一個(gè)OFDM符號(hào)的平均信號(hào)功率的歸一化。對(duì)于DFTS-OFD

18、M,PRPA明顯降低，相比OFDM,再考慮到移動(dòng)終端的電源能力，這種傳輸技術(shù)在上行鏈路的傳輸中是非常有用的。</p><p>　　DFTS-OFDM信號(hào)解調(diào)的基本原理如圖2-9所示，“DFT的OFDM解調(diào)”。這些操作和圖2-9“DFT的OFDM解調(diào)”基本上是相反的。即size-n離散傅里葉變換處理中，和接受信號(hào)不對(duì)應(yīng)的頻率采樣會(huì)被移除。</p><p>　　圖2-9 DFTS OFDM調(diào)制

19、</p><p>　　LTE multiple access techniques</p><p>　　LTE multiple access</p><p>　　OFDM transmission</p><p>　　Orthogonal Frequency Division Multiplexing (OFDM) is a multica

20、rrier transmission</p><p>　　technique that has been adopted as the downlink transmission scheme for the 3GPP</p><p>　　Long-Term Evolution (LTE) and is also used for several other radio technolog

21、ies, e.g.</p><p>　　WiMAX and the DVB broadcast technologies.</p><p>　　It is characterized by a tight frequency-domain packing of the subcarriers with a subcarrier spacing f = 1/Tu, where Tu is t

22、he per-subcarrier modulation-symbol time. (See Figure 2-1, “OFDM subcarrier spacing”) .</p><p>　　OFDM transmission is block-based. During each OFDM symbol interval, modulation</p><p>　　symbols a

23、re transmitted in parallel. The modulation symbols can be from any modulation alphabet, such as QPSK, 16QAM, or 64QAM.</p><p>　　For 3GPP LTE, the basic subcarrier spacing equals 15 kHz. On the other hand, th

24、e</p><p>　　number of subcarriers depends on the transmission bandwidth, with in the order of 600 subcarriers in case of operation in a 10 MHz spectrum allocation and correspondingly fewer/more subcarriers in

25、 case of smaller/larger overall transmission bandwidths.</p><p>　　Figure 2-1 OFDM subcarrier spacing</p><p>　　The physical resource in case of OFDM transmission is often illustrated as a</p&g

26、t;<p>　　time-frequency grid where a column corresponds to one OFDM symbol (time) and a row corresponds to one OFDM subcarrier, as illustrated in (see Figure 2-2, “OFDM time-frequency grid” ).</p><p>　

27、　In the ideal case, despite the fact that the spectrum of neighbor subcarriers do overlap, the OFDM subcarriers do not cause any interference to each other after demodulation due to the specific choice of a subcarrier sp

28、acing f equal to the modulation symbol rate.</p><p>　　Figure 2-2 OFDM time-frequency grid</p><p>　　An OFDM signal sampled at a rate fs = N × f is the size-N Inverse Discrete Fourier</p&g

29、t;<p>　　Transform (IDFT) of the block of modulation symbols a0, a1,...aN-1. Thus, OFDM</p><p>　　modulation can be implemented by means of IDFT processing followed by</p><p>　　digital-to-a

30、nalog conversion (see Figure 2-3, “OFDM modulation”) . In practice,the OFDM modulation can be implemented by means of Inverse Fast Fourier Transform (IFFT) easy and fast processing, by selecting the IDFT size N equal to

31、2m for some integerm. At the receiver, by sampling the received signal at the rate fs = N× f, efficient FFT processing is used to achieve OFDM demodulation and retrieve the block of modulation symbols a0, a1,...aN-1

32、( see Figure 2-4, “OFDM demodulation”) .</p><p>　　Figure 2-3 OFDM modulation</p><p>　　Figure 2-4 OFDM demodulation</p><p>　　As mentioned above, an uncorrupted OFDM signal can be dem

33、odulated without any</p><p>　　interference between subcarriers. However, in case of a time-dispersive channel (such as multipath radio channels), the orthogonality between the subcarriers is lost, causing In

34、ter Symbol Interference (ISI). The reason for this is that the demodulator correlation interval for one path will overlap with the symbol boundary of a different path (see Figure 2-5,“Time dispersion and corresponding re

35、ceived signal”) </p><p>　　Figure 2-5 Time dispersion and corresponding received signal</p><p>　　To deal with this problem and make an OFDM signal truly insensitive to time dispersion on the radi

36、o channel, so-called Cyclic Prefix insertion is typically used in case of OFDM transmission. As illustrated in(see Figure 2-6, “Cyclic Prefix insertion”) , cyclic-prefix insertion implies that the last part of the OFDM s

37、ymbol (the last Ncp symbols) is copied and inserted at the beginning of the OFDM block, increasing thus the length of the OFDM symbol from Tu to Tu + Tcp, where Tcp = Ncp,Tu is the l</p><p>　　The OFDM symbol

38、 rate as is reduced as a consequence. Thus, subcarrier orthogonality is preserved in case of a time-dispersive channel, as long as the span of the time dispersion is shorter than the cyclic-prefix length.</p><

39、p>　　Figure 2-6 Cyclic Prefix insertion</p><p>　　The drawback of cyclic-prefix insertion is that it implies a corresponding loss in terms of throughput as the OFDM symbol rate is reduced without a correspo

40、nding reduction in the overall signal bandwidth.</p><p>　　The combination of OFDM modulation (IFFT processing), a (time-dispersive) radio</p><p>　　channel, and OFDM demodulation (FFT processing)

41、 can then be seen as a</p><p>　　frequency-domain channel as illustrated in(see Figure 2-7, “Frequency domain model of OFDM transmission reception”) , where during each OFDM symbol time period, N different mo

42、dulation symbols are transmitted, each on a given subcarrier over the corresponding sub-band, in contrast to single wideband carrier systems, such as a WCDMA where each modulation symbol is transmitted over the entire ba

43、ndwidth.</p><p>　　Figure 2-7 Frequency domain model of OFDM transmission reception</p><p>　　On frequency channel k, modulation symbol ak is scaled and phase rotated by the</p><p>　　

44、complex (frequency-domain) channel coefficient Hk. At the receiver side, to allow for</p><p>　　proper decoding of the transmitted information after demodulation, the receiver needs an estimate of the frequen

45、cy-domain channel taps H0, H1,...,HN-1. This can be done by inserting known reference symbols, sometimes also referred to as pilot symbols or pilots,at regular intervals within the OFDM time/frequency grid. Using knowled

46、ge about the reference symbols, the receiver can estimate the (frequency-domain) channel taps necessary for the decoding.</p><p>　　OFDM signal bandwidth</p><p>　　The basic bandwidth of an OFDM s

47、ignal equals N × f, i.e. the number of subcarriers</p><p>　　multiplied by the subcarrier spacing. On the other hand, by setting the symbols to be</p><p>　　transmitted on a group of side con

48、tiguous subcarriers to zero, the basic bandwidth is</p><p>　　reduced to Nc × f where Nc is the number of non-null subcarriers. However, the spectrum of an OFDM signal falls off slowly outside the basic

49、OFDM bandwidth and especially much slower than for a WCDMA signal. Thus, in practice, typically in the order of 10% guard-band is needed for an OFDM signal, implying that, as an example, in a spectrum allocation of 5 MHz

50、, the basic OFDM bandwidth Nc × f could be in the order of 4.5 MHz. Assuming, for example, a subcarrier spacing of 15 kHz as selected for L</p><p>　　DFTS OFDM transmission</p><p>　　Discrete

51、 Fourier Transform Spread OFDM (DFTS-OFDM) is a transmission scheme that has been selected as the uplink transmission scheme for LTE. The basic principle of DFTS-OFDM transmission is illustrated in(see Figure 2-8, “DFTS

52、OFDM signal generation”) . Similar to OFDM modulation, DFTS-OFDM relies on block-based signal generation. In case of DFTS-OFDM, a block of M modulation symbols from some modulation alphabet, e.g. QPSK or 16QAM, is first

53、applied to a size-M DFT. The output of the DFT is th</p><p>　　Figure 2-8 DFTS OFDM signal generation</p><p>　　Comparing (see Figure 2-8, “DFTS OFDM signal generation” ),with the IFFT-based imple

54、mentation of OFDM modulation, it is obvious that DFTS-OFDM can alternatively be seen as OFDM modulation preceded by a DFT operation.</p><p>　　If the DFT size M equals the IDFT size N, the cascaded DFT and ID

55、FT blocks of (see Figure 2-8, “DFTS OFDM signal generation”), will completely cancel out each other.</p><p>　　However, if M is smaller than N and the remaining inputs to the IDFT are set to zero, the output

56、of the IDFT will be a signal with low power variations, similar to a single-carrier signal. Besides, by varying the block size M the instantaneous bandwidth of the transmitted signal can be varied, allowing for flexible-

57、bandwidth assignment. The main benefit of DFTS-OFDM, compared to a multi-carrier transmission scheme such as OFDM, is reduced variations in the instantaneous transmit power, implying </p><p>　　The basic prin

58、ciple of DFTS-OFDM signal demodulation is illustrated in (see Figure 2-9,“DFTS OFDM demodulation”) . The operations are basically the reverse of those for the DFTS-OFDM signal generation of (see Figure 2-9, “DFTS OFDM de

59、modulation”) , i.e. size-N DFT (FFT) processing, removal of the frequency samples not corresponding to the signal to be received, and size-M Inverse DFT processing.</p><p>　　Figure 2-9 DFTS OFDM demodulation